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A collection of ferrite toroid related hobby research spin-offs |
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Some handy stuff for the occasional ferrite user |
© PROJECT ENGINEER: WALTER - PE1ABR  Zie je liever een Nederlandse versie?? Klik op de vlag of HIER |
| The pictures are all clickable
for larger versions and/or PDF Ferrite What kind of stuff is it?? How is it made?? 
↑ Overview PDF 
↑ 3C11 - WHITE PDF 
↑ 3E25 - ORANGE PDF 
↑ 3F3 - DARKBLUE PDF 
↑ 4A11 - PINK PDF 
↑ 4C6 of 4C65 - VIOLET |
Just like iron (or better : steel) is a practical magnetic material for low frequencies (f.i. in mains transformers), for higher frequencies ferrite is a handy and better material.
Ferrite "to the bone" is nothing else than a ceramic crystal shape of iron-oxide - a kind of baked rust......
Unfortunately the electrical conduction is far too high, so the losses too much.
With a trick science found a solution, that is they deliberately polluted the crystal form with other oxide-materials that reduced conductivity between the tiny local microcrystal areas. And at the same time try to keep the magnetic properties.
Dependent on the kind of additions, the magnetic properties become gradually less and the electrical resistance higher and higher. And the usable frequency also gets gradually higher. See the upper picture left with some (ex-)Philips types, more and more a bit lower and at the same time higher in frequency. What you don't see are curves of the losses. Those are the dashed lines in the individual material curves. More about them later. About the oxide-mixtures that are added: two different kind of "tastes" have evolved. The main types: or MnZnFe(II)- , or NiZnFe(II)-oxide mix with lots of Fe2O3 iron(III) -oxide (= ferrite). The main components after ferrite itself are either manganese-zinc (MnZn) or nickel-zinc (NiZn) oxide. And some more other metal oxides, a.o. vanadium Va, copper Cu, magnesium Mg, cobalt Co, tin Sn, and probably many other secret materials. After a double sintering (annealing) action cubic spinel crystals (20 to 30 micrometers) arise with still reasonable conductivity, but isolated from each other by the remaining mixtures between the crystals (typically ZnO - zinc oxide). Manganese-zinc as mix for ferrite gives a still noticeable (with an ohm meter measurable) conductivity and rather high magnetic properties. Nickel-zinc as addition reduces the magnetic properties much more and gives a very high (almost immeasurable) electrical resistance. Higher frequencies demand for very low losses due to the higher number of magnetisations per second. And a too high magnetic property is also undesirable, so all OK, each variation of "taste" has its own use. Another word to avoid confusion between two magnetic concepts: The ferrite material constant at extremely small field strengths in a closed system, whatever its size, is called the mu (µ). This can be seen as the "gain" compared to air. Also called the µi, the initial permeability. This is the material property that you see everywhere in the tables, it is independent of the core design or shape. The relative self-induction of one specific core, with all data in it, a parameter of core thickness, core length and the typical mu of the core material is called the AL or AL value. Note: For cores between 30 and 40 millimeters, depending on the thickness, the conversion factor of the length / surface average and AL is around the factor "1". Make sure you take the AL, which belongs to the core and not the mu! The magnetic conductivity properties combined with the shape (square diameter and length) properties can be expressed in a practical standard value for each make and type of toroid: the AL value, or better written down: AL value. It expresses the number of nanoHenries per turn, or millihenries per 1000 turns. For both expressions it stays the same AL value "number". One more time: The ferrite property independent of the shape is called the mu (µ). The optimal range of use for MnZn- and NiZn-ferrite have some overlap, where MnZn is usable from LF to 0.25 - 2 MHz, NiZn can be used from 1 MHz to far beyond 100 MHz for energy transfer (= transmitters). With limitations it is even used up to 1 GHz. It all can be seen nicely in the upper picture where the magnetic property mu (µ) for a number of materials is plotted against the frequency. NiZn mu (µ) between 100 and 750 and MnZn from 1000 to 15000. The average AL value (depends on geometric properties) goes for "average" toroids between 1000 - 15000 for MnZn and for NiZn from about some dozens to about 1000. Average is a toroid between 10 mm and 36 mm. In this Philips picture MnZn is the type starting with a "3" and NiZn is the material type starting with a "4". For EMC and suppression applications the loss factor is an enhancing factor, it is added to the inductance Z. In this case they form together a complex impedance (Ztot = ZXL + jR ). In EMC applications there is NO energy transfer, energy should be absorbed! What you can say about mains oriented electronics for coils and resistors, you can speak about real (R) and imaginary (L) impedances, you can also say about EMC ferrites: you have a lossless SELF-INDUCTANCE part (ZXL) and a complex LOSSY part (jR). But compared to mains electronics real and imaginary are somewhat "twisted" in their meaning.....! Just to emphasize: No ferrite has a stable fixed value mu. All ferrites have a frequency at which the inductance effect collapses. So they have a typical "terminal value". Click to see the Philips collection image (also top-left). At the bend in the curve losses rise exponentially! For "power applications" preferably as late as possible a rising "dotted" graph. Especially for EMC application this part of the dotted curve is deliberately increased and given a broad back. For HF transformer applications you'd rather NOT use this type. Material which has been made to obtain the opposite effect - as tight as possible dotted curve - is called "low-loss" in the Cros_Ref_list. Except ferrite, microscopic fine iron powder is also used as toroid material in combination with a clay (becomes → ceramic) binder. Low frequency power applications (switch-PSU ripple chokes) or cheap mains-filters (light-dimmers) mostly use "standard" iron powder. There are powder iron cores available with only low frequency application (hydrogen reduced Fe-oxide), depending on type only from 10 kHz to 1 MHz usable. Usually for power supplies and filtering. The only advantage compared to ferrite is a to 4x higher saturation. For those low frequencies no special HF loss demands. So this kind of iron powder toroid is useless for HF purposes! The iron powder for HF application cores is chemically made by reducing iron-containing chemicals (Fe-pentacarbonyl) with the "charcoal" principle. It gives microscopic fine isolated (foam-like) iron dust. Almost all the Amidon iron powderdust toroids are of this kind. Keep in mind: compared to ferrite this material has an extreme low magnetic property. Amidon powder materials µ is in the range: between 5 and 30 !! So the average AL value for 25 to 40 mm cores is also between 5 and 30 (nH/turn or mH/1000 turns). When expressed in the same order of magnitude (units) as ferrite! = nH / winding or mH/1000 windings. Posted in the Philips collection image chart top-left it is on the "zero" line or one indent above. To make it prone for error: due to these low values for iron powder cores the industry has chosen a different standard!! It is a factor 10 higher and overcomes the need to use values with a decimal point in it. They therefor express the AL value in microHenry (µH) per 100 turns!! If you are going to use (my) standard formulae take notice of the fact that you have to divide ALL Amidon values by 10 to use the same calculating method!! |
How do I make the right ferrite choice?? Try it 100 times??? 
How does it behave with (power) losses?? 
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The ferrite or iron powder choice is demanded mostly by the application and circuit impedances. If you use ferrites for resonance circuits (high impedances) you have to be careful to make a choice or you run into trouble!! Already at low frequencies the circuit damping by ferrite losses is such that the max reachable Q is seriously spoiled by the loss factor. A resonance circuit is NOT a low-ohmic transformer application, but a tuned LC-circuit. The loss characteristics are the dashed lines in the ferrite pictures above. If the value rises above 5 to 10% of the non-dashed curve, the material is less-ideal for resonant LC applications. At 25% of the value worthless! NOT the LC and the load dictates the Q, but the losses then do! During experiments I drew my own limits at about 7.5%. An example of the resulting limits for resonance applications: 4C65 = ±15MHz, 4A11 = ±700KHz, 3F3 = ±400KHz, 3C85 = ±200KHz, 3C11 = ±100KHz en 3E25 = ±80KHz !!! The loss factor has also influence in normal "voltage" transformers at power applications. Heat !! If impedances are low and you make a good NiZn choice, all goes well. (the range 25 - 500 ohm is commonly used...) Used in applications without power and again: low impedances ( like in Rx and data transfer = ethernet), MnZn is not such a problem. You only have to take into account that there could be a parasitic ballast-R due to losses. Used in current (transmission line) transformers ( = SWR suppressor, see further down) the main current is no problem, it cancels out! The common mode current that must be suppressed is now demanding. Because this current is suppressed and partly by the application (eg symmetric to asymmetric and excessive impedances after the faulty use of a tuner), it creates a high voltage across the inductance of the cable on the ringcore. This voltage is therefore generated by the flux-changes in the core as back-EMF in order to suppress this current. If the external voltage becomes too high or too short of core material surface area is used (the flux-change does not reach the required back-EMF voltage), then the core is saturated and gets burning hot due to power absorbtion. Either the windings flies into fire, or the core disassembles with a bang! The only method that works is much more core surface area (divide fluxenergy over more cores and / or winding units, see picture) and also no false application were too high impedances may occur. This increases the voltage strongly and the problems! Voltage according to P = U2 / R , play with 10, 100, 1000 W and 5 (semiconductor amplifier), 50, 500 and 5000 Ohm!! But for resonance circuits iron powder cores are mostly a better choice. Dependent of the frequency range only the better NiZn is only usable. It is always a compromise between the number of turns and the material choice. The higher the AL value the worse for resonance on higher frequencies. And the problem with the lower AL value: a unrealistic number of turns. So stacking toroids could be a solution. |
| How many windings are needed for a certain inductance??
Calculate AL with the f-res test 
My own Software for above!! FERRICALC Postscript overlay sheet |
Make a ferrite choice with the minimum frequency in mind. Do NOT go as low as possible, but try to find what is usable with a very limited number (5 to 8) of turns for the Zmin. If you have measured them yourself, this is probably the material with the best Q. In the links further down you can find some "extracts" (in DUTCH) from my booklet with calculation tricks to calculate fmin at N=x (x = 5 to 8) for a number of toroids that you have in stock.
For 50 Ohm circuits also choose Zmin = 4 times 50 = 200 Ohm as minimum coil impedance. Use the AL value that belongs to that toroid (in a manufacturers table or in your home made ones), use the appropriate formulae N = 1000.√(L/AL) and finished. Get my AL table extract (in DUTCH) from "Ferrite Info" in PDF, with also extra explanation. There's a lack of space over here. Some further down another extract about measuring is spoken off. Get it now already. Click here But how do you get those AL values from unknown toroids?? Again: further down.... |
How do I make measurements??
Measuring with the TOP-R method
Or with the TOP-C method
My handy measuring cable in Elektor!
The first older modules
The impoved rebuild modules
My own calculation Software!! switchable between NL and the UK 
FERRICALC
The results are patched as an EPS print-overlay in the original (renamed) PS print file. This should be available in the same folder. So also pick up the postscript sheet. 
↑ An example to fill-in in PDF programmed 2x on 1 A4 sheet (1x = equal to the EPS overlay) |
Standard ferrite properties (a.o. the µ and AL value) are only measured reliable and simple without too much parasitic effects on rather low frequencies.
Because also the (too huge) number of windings has a parasitic effect, a nice agreement is: 10 windings divided over the total circumference, no more and no less. 10 = the number of turns through the centre hole, NOT the turns visible on the outside!! Think about a current clamp → 1 pass = 1 total winding!
Also keep this in mind: only with exactly 10 windings on a toroid you can have a direct fast impression of the AL value on a coil (L-)meter. Because only for 10 windings is valid: AL value = shown µH value x 10. Which works only if the coil meter measures on a low frequency before the decay in magnetic properties!!!! Measuring the AL of most unknown ferrites with a dipper (or MFJ device or RF-Analyst) gives NONSENSE results! See article in Elektor written by PE1ABR for more background information about direct AL readout. Also published in the UK (and Germany, France, the Netherlands, etc.). To reduce parasitic toroid and C effects a first measurement with at least 1 to 2 nF (1%) in parallel is OK. Otherwise the resonant frequency is too high. A second (additional) measure, for material with a relative high AL value (above 2000 - 3500), is a second measure with 10 nF (1%) in parallel. The values for AL shouldn't differ too much, although the Q mostly is more advantageous. The used capacitor values should be verified on a RCL bridge or aiding-oscillator circuit. Also keep in mind to add the scope probe capacitance!! See further down. Another measurement can be added for higher AL material: for a better "total" view test it also with 25 windings and the same 1 nF capacitor. With or a 10x bigger C and/or a larger winding number you get more "down" to the smoother part of the AL (or better : mu) curve, see the top picture for an overview. Some ferrites (high AL) or mains filter toroids can better be measured with even a larger capacitor than 10 nF, f.i. a 100 nF is OK. You are measuring pure low frequency (LF = audio range) than. Very low capacitance values (under 1 nF) gives more unwanted parasitic effects and make the measurement completely unusable. As well as with a low-ohmic current link as well as a high impedance voltage link errors may occur. Mostly in the Q value, hardly in the AL. Some sort of high impedance series link is also used by the industry (Murata, for their IF filter coils), that method appeared handy. On the other hand, a current link, feeding a link from an impedance under 10 Ohms, looks more like a short circuit, so unusable in my opinion to make a reliable undamped (!!) Q measurement in this circuit. An oscilloscope 10 MegaOhm probe is in parallel (with its capacitance from about 15 - 25 pF, → add it !!) to the LC circuit. As a feeding control signal we use a simple sine generator from LF to about 2 - 5 MHz. For very small NiZn toroids (under 10 mm) this is sometimes not high enough. Better use a higher parallel capacitor instead to reduce the resonant frequency. At the end of the feeding coax-cable it is advicable to add a 50 Ohm terminator R, and from this point the high feeding R to the LC measuring circuit. For the ease of use, and if not too long cables are used, it is possible to misuse an uncompensated 1x scope probe as feeding cable to the series R's. (a probe without any R or C correction and used backwards). For the ease of use it has a nice clip to attach, see picture. Via this series R, a choice from the values 1K, 10K, 50K and 100K Ohm (or more..., up to 500 kOhm..), the signal goes to the LC circuit. The f-res is found by quickly tune the whole frequency range of the generator, under use of the lowest series R. Otherwise you could "fly over it", it could be difficult to find. After that use the highest possible value. You have the least degradation in Q due to the highest resistor. Although a scope is the most easiest to work with as an indicator, an amplified HF millivoltmeter also could work. It all depends on the max generator voltage, sensitivity of the scope/millivoltmeter and also the losses in the ferrite. After reading technical papers from others (eg. ON9CVD), is measuring, NOT with a high feed-R, but with a very small feed-C, also a very good method. I have added jumper selectable capacitor steps between 1 and 0.1% of the reference C. Also, not measuring with two, but with THREE versions of the test module with 1 nF, 10 nF and 100 nF gives a better "overall" view of the effect of the "inclined plane" of the mu curve. Most worst ferrite (MnZn with the highest AL and / or the lowest "kink") did indeed gave only with the 100 nF Cap. realistic Q measurement results. The once-selling blue 23 mm Philips toroids appeared useless just above 20 kHz!!!!!!!!!!! If you want to calculate accurately after using the TOP-C modules, for each branch-C the corresponding different ref-C value must applied. That's again a slight disadvantage. The main advantage is that you no longer need to take the supply-R damping into account. Look at the TOP-R drawing or the TOP-C drawing in PDF. From these measuring aids on a piece of clad board you need at least THREE, more even better! also see both IMAGES left. The new modules in the picture are made slightly different than in the drawing. The drawing was already finished ...... Principle is the exactly the same, but the typical changes are untidy on the drawing. In the picture the jumpers are placed between the Cref-trace and the four C-tap's. Each C-tap capacitor has here its own 50 ohm dummy-R. Because of this the other C-tap's that are not used form no parasitic load present at C-Cref. As shown in the drawing: only one time a 50 ohm dummy, a set of jumpers, and then C-tap's, all connected to the Cref as shown, is also possible. Etching is not needed, a sharp knife or dental milling tool works fine (need for plasters??) To reduce parasitics you should remove unneeded copper. QB√2 The ferrite quality (max. usable f and losses at f-res) can be judged by comparing the Q at f-res from different measured toroids. Or with an identical measurement on a different Cref module. The Q follows out of the two 70% points f1 and f2 next to f-res (±70% = -3dB = B-sq.root-2 = B√2) and put in the formulae ( QB√2 = f-res/(f1-f2) ) This can be corrected - series R damping effect removed - to calculate the real Q without the feeding R. This correction depends strongly on parallel C and frequency. You need a correction graph for every measuring module, and also every series R variation, you make. To encourage everyone to take toroid measurements, I have put all the formulas in a Windows program and made it a little "idiot-proof". That's easy! For every toroid you intensely measured you can keep a "datasheet" page in a small A5 binder. But start with a few well-known toroids to get familiar..... Much more additional information about the measuring method, formulae and Q-correction tricks in the following piece taken from the booklet "Ferriet Info" IN THE DUTCH LANGUAGE!! Click over here for the DUTCH document in PDF |
↑ Example filled in by hand in a PDF |
Click icon to get VB6 runtime and OCX files for FERRICALC if needed. |
↑ 1 left-over empty line patched with the EPS overlay (PDF) |
How should it be winded?? Current and voltage transfer 
The strong negative impact of the para-C already at low winding numbers. 
A Guanella wideband 50 ohm in/out common mode suppression filter 
↑ PDF
↑ PDF
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How to wind?? Depends on the application, is it a current transformer or voltage transformer. A "current" transformer usually consists of one or more transmission lines between in- and output. The international name is "transmission line transformer". Common mode currents are suppressed by the choke effect. The (bifilar) winding wires form, just like HF-cabling, a fixed constant impedance on the cores. The parasitic capacitance is part of the cabling strains impedance and doesn't harm. The capacity to or through the core is a pain. By adding more branches, in parallel on one side and in series at the other, an impedance transfer is accomplished. Those windings made with double (bifilar) wire are extremely tightly twisted, sometimes even in (teflon) sleeves. The mechanical construction of the strains is such that they form the ideal impedance to sum or divide to the right value on each side to match the connected impedance. An example: Two strains of 100 Ohm cabling on the cores form 50 Ohms parallel on one side and 2 times 100 Ohm = 200 Ohms in series at the other side. The common mode choke value must be a few times the connected impedance. Up to 10x the impedance, also with a low parasitic C between input and output. Because the Z of a single toroid mostly is not enough, (many) more toroids are stacked. And see a little higher in the text, note the maximum voltage that can occur and that the Z is stable low. If problems ==> saturation and fire. (This is called the Guanella transmission line principle) The other method is the "standard" transformer = a voltage converter unit, the famous MLB. OK, that's slightly harder to get it working. The problems on the high side A wideband (voltage) transformer also works best with LOW impedances, it suffers more from parasitic capacitance. Each of the windings must cover the total circumference of the toroid. Leave some clearance between begin and end of each winding due to the parasitic capacitance. Keep in mind: an autotransformer with a tap has less wideband characteristics than a version where each winding covers the total circumference!! By a less ideal coverage of the windings over the circumference there is a less ideal transfer from one winding to the other. This results in "leakage inductance", a parasitic series inductance. At the point where those parasitics are noticeable at the high end the output voltage drops (sagged dashed line). At the "low" side the lack of inductance causes the drop. Both effects can be compensated somewhat by adding capacitors in parallel or series. The parallel capacitor to compensate the high-end drop (and lift the sag) should be connected at the secundary side. Secundair is seen from the origin of the HF!! It only works to compensate the parasitic series inductance (= leak field compensation). Is the parasitic parallel capacitor already too high?? Than it no longer works... You can measure the leak field inductance by making a short circuit at one winding (high side). What is left over from the transformer inductance can be measured at the other winding (50 ohm side): that's the leak field parasitic inductance. The problems on the low side Used as aerial transformer there are hardly any problems due to the declining Z. As interstage transformer in an amplifier, there is a possibility for parasitic oscillations by the declining of the total impedance. The Z of the series capacitor compensates for the drop in inductance at the low side at the point that the inductance Z is becoming too low to match to 50 Ohm, so the winding becomes a HF short circuit. ( C = L/R2 ) Tho test the effect of the large series or small parallel capacitor on an analyzer in an empirical setting, you can do that with a homemade capacitor decade bench. To obtain a low residual capacitance and in one setting a lot of values I do this with a mini hexadecimal (ABCD-)switch on a piece of experiment board. Sometimes with an extra on/off switch added. On it 4 (or 5) sets of C's in ascending values in the ratio 1-2-4-8(-16). You get rather easy 15 or 31-step linear incremental values (and step 16/32 = the value zero = No C). Extra additional information on the MLB application While there is still some doubt by some, a smart chosen set of two different toroids does have some advantages over a single toroid! If you only use a (common) low AL toroid, mostly NiZn, transfer is ideal only over 4 MHz! Or you should add a lot more windings, but in that case 30MHz could already be a problem. If you only use a toroid with a much higher AL value, mostly MnZn, the transfer starts nice low (longwave, mediumwave or even under 100kHz on VLF!), but transfer is useless above 1 to 2 MHz. For receiving-only purposes a duo-toroid does has advantages! Hundreds of measurements have shown this. By using a MnZn toroid for the lowest range you can limit the number of turns considerably. Because it is so low it is advicable to use double wire (against leak field!!) and NO copper foil strains. This is a huge advantage to limit the interwinding capacitance. By that the transfer is still OK on the highest frequency! Over there the transfer is done by the NiZn material, but the MnZn causes some absorption. Besides the loss in the MnZn core there is unfortunately also a larger leak inductance due to the decline of the AL in the MnZn core. That's fine to compensate with a small additional C of about 20 pF. The diagram, as can be found in the column left, with separate windings also gives an extreme advantage over all commercial available products. EVERYTHING on the market does not use separate masses. And the wrong twist winding method with a too high para-C or the winding tap / auto transformer method. Between 15 and 20 MHz there may already be more capacitive than inductive transfer! Sometimes not even an extra ground wire connection to the antenna / coax transition. I.e. the coaxial cable (sheath) is ALWAYS the HF counterpart, up to inside-home near your equipment. An additional current transformer to suppress man-made noise (blocking aerial cable reception) also blocks the HF counterpart. With my MLB version all is NO problem. You can inter-connect both earth's or not. You MUST always connect a RF counterpart. And finally it MUST be connected to outside ground somewhere. This is also the proper static discharge clearing. Only in this way a current transformer is perfectly complementary. Or better use two, one near the aerial and one near the cable entrance in your home. Stubborn? It might help by telling that the effect of all possible winding methods is measured and tested 100-fold ..... Only at low (antenna) impedances and low power the absorbing impact effects of the MnZn ring on the power or signal is negligible. You can compare it with a parasitic "rear light of a bicycle" which is in parallel to the bicycle headlamp. It affects the headlamp light, but one can live with it! If you use this duo-system to feed a NON QRP transmitter to an aerial you could have a problem. The MnZn toroid could become HOT!! |
What has come out of it....?????  |
After two periods of measurements and thoughts all my efforts have had the result that a Dutch language syllabus has been made with all the results bundled.
The first period of a few years was mainly exploring and gave me the insight what matters, how to measure, where to take account off, and so on. The first set of measurements was hereby unreliable and probably worthless.... In a second period of measurements all was better understood and better documented. What is the most ideal measurement method, what is the best winding method, which type of toroids are most ideal to make combinations. All with the idea in mind to make the most ideal longwire aerial balun transformer. With predefined start frequency and predictable frequency working range. It all had as a result the publication of the Dutch language booklet "FERRIET INFO" (= "Ferrite Info"). Many, many hundreds of them have been sold by VERON (the Dutch RSGB / ARRL organisation). The original work didn't account for the parasitic effects at resonance and power use. It was initially not meant to cope with that. It is felt nowadays as a lack of information, so extra info is added ( or must still be added....) to the PDF's. On this web-page you already can read a preview of the headlines. Also in the second part with lots of practical designs and the use of ferrites for Rx purposes are updated to the situation 14 years later. A.o. an active aerial with a very huge intercept point due to a special circuit trick. With recent layouts of the circuit boards and setup drawings in color! {Because both the total quiescent current and the division between two JFET's in a semi-balanced stageset is adjustable, there is a point at which the nonlinear properties of one JFET in severe overload can be offset by exactly the mirror image of the other JFET . Thereby you turn the mixing products (f1 + f2 = f3) way to ZERO! An additional trick are the cascode JFET stages with a mini-power transistor in the drain, causing almost NO drain voltage variation. As a result, the NON-linear effect of the Miller capacity also is gone!} Part three, the EMC part with lots of interference experience, is somewhat outdated... or isn't it?? We now have the CE commitment. Problems are still the same, but with other type of equipment (PLC, a.s.o.)?? Over here some slow FTP links to my Homeserver where the three parts can be downloaded as PDF (IN DUTCH !!)
TEST Ferrite Info Booklet part 1 link (IN DUTCH !!)
TEST Ferrite Info Booklet part 2 link (IN DUTCH !!)
TEST Ferrite Info Booklet part 3 link (IN DUTCH !!)
There is still work to be done to update them. That is, they are not final. |
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Back to the Ferrite page with manufacturers or Back to the Inductor page with suppliers and components or Back to Quick menu at the HOME page or to the top of |
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HOMESERVER TEST FTP link Tries to fetch a dummy from the Steward Company |
| ferriet boekje info -- ferrietUK.htm by Walter - PE1ABR - 2011-10-21 |